Distributed photodiode structure having majority dopant gradient and method for making same

ABSTRACT

A distributed photodiode structure is shown formed on a semiconductor substrate having a first dopant type where a first plurality of diffusions of a second dopant type are formed on a first surface of the substrate. A second plurality of diffusions having the first dopant type are formed on the first surface of the substrate between the first plurality of diffusions. In a further refinement, a second surface of the substrate is diffused with the first dopant type. In yet another refinement, a plurality of trenches are formed on the first surface and the second plurality of diffusions are formed within the trenches.

This application is a continuation-in-part of U.S. application Ser. No.09/448,861, filed Nov. 23, 1999 now U.S. Pat. No. 6,548,878, which is acontinuation-in-part of U.S. application Ser. No. 09/037,258, filed Mar.9, 1998, now U.S. Pat. No. 6,198,118 B1, which issued Mar. 6, 2001. Thisapplication further claims the benefit of U.S. Provisional ApplicationNo. 60/222,296, filed Aug. 1, 2000.

BACKGROUND OF THE INVENTION

1. Field of Invention

The present invention relates to a photodiode which can be fabricatedusing standard semiconductor fabrication techniques.

2. Description of the Related Art

Infrared wireless data communication is a useful method for short range(in the approximate range of 0-10 meters) wireless transfer of databetween electronic equipment; such as, cellular phones, computers,computer peripherals (printers, modems, keyboards, cursor controldevices, etc.), electronic keys, electronic ID devices, and networkequipment. Infrared wireless communication devices typically have theadvantages of smaller size, lower cost, fewer regulatory requirements,and a well defined transmission coverage area as compared to radiofrequency wireless technology (i.e. the zone of transmission is boundedby physical walls and therefore more useful in an office environment).In addition, infrared wireless communication has further advantages withregard to reliability, electromagnetic compatibility, multiplexingcapability, easier mechanical design, and convenience to the user ascompared to cable based communication technology. As a result, infrareddata communication devices are useful for replacing 0-10 meter long datatransfer cables between electronic devices, provided that their size andcosts can be reduced to that of comparable cable technology. As examplesof the type of wireless communications links that are presently in use,the Infrared Data Association (IrDA) Physical Layer Link Specification1.le specifies two main physical layer infrared modulation protocols.

The IrDA Physical Layer Link Specification 1.le also specifies two modesfor modulation of data on the infrared transmitted signal. One mode is alow-speed (2 Kbp/s to 1.15 Mbp/s) on-off infrared carrier usingsynchronous modulation where the presence of a pulse indicates a 0 bitand the absence of a pulse indicates a 1 bit. The second mode is a highspeed (4 Mb/s) synchronous Four Pulse Position Modulation (4 PPM) methodin which the time position of a 125 nS infrared pulse in a 500 nS frameencodes two bits of information. The 1.le specification also specifies apreamble pattern which is sixteen repeated transmissions of apredetermined set of symbols.

Infrared data communications devices typically consist of transmitterand receiver components. The infrared data transmitter section consistsof one or more infrared light emitting diodes (LEDs), an infrared lens,and an LED current driver. A conventional infrared data receivertypically consists of an infrared photodiode and a high gain receiveramplifier with various signal processing functions, such as automaticgain control (AGC), background current cancelling, filtering, anddemodulation. For one-directional data transfer, only a transmitter atthe originating end and a receiver at the answering end is required. Forbi-directional communication, a receiver and transmitter at each end isrequired. A combined transmitter and receiver is called a transceiver.

A representative example of a conventional infrared data transmitter andreceiver pair is shown in FIG. 1A. Infrared transmitter 10 includes LED16 which generates a modulated infrared pulse in response to transistor14 being driven by the data signal input at D_(IR),. The modulatedinfrared signal is optically coupled to an infrared detector, such asphotodiode 24 normally operated in current mode (versus voltage mode)producing an output current which is a linear analog of the opticalinfrared signal falling on it. The infrared pulses generated by LED 16strike photodiode 24 causing it to conduct current responsive to thedata signal input at D_(IR) thereby generating a data signal received atD_(IR).

In receiver 20, the signal received at D_(IR), is transformed into avoltage signal V_(IR) and amplified by amplifier 26. The signal outputfrom amplifier 26 then feeds into comparator 42 which demodulates thereceived signal by comparing it to a detection threshold voltage V_(DET)in order to produce a digital output data signal at D_(OUT). Thereceived signal waveform will have edges with slope and will ofteninclude a superimposed noise signal. As a result, V_(DET) is ideallyplaced at the center of the received signal waveform so that the outputdata signal has a consistent waveform width despite the slope of thereceived signal edges. Also, placing V_(DET) the center of the receivedsignal -improves the noise immunity of receiver 20 because the voltagedifference between V_(DET) and both the high and low levels of thereceived signal is maximized such that noise peaks are less likely toresult in spurious transitions in D_(OUT).

The received signal, however, can vary in amplitude by several orders ofmagnitude due primarily to variations in the distance betweentransmitter 10 and receiver 20. The strength of the received signaldecreases proportional to the square of the distance. Depending on therange and intensity of the infrared transmitter, the photodiode outputssignal current in the range of 5 nA to 5 mA plus DC and AC currentsarising from ambient infrared sources of sunlight, incandescent andfluorescent lighting. As a consequence, the center of the receivedsignal waveform will vary, whereas V_(DET) Must generally be maintainedat a constant level. To address this problem, receivers typicallyinclude an automatic gain control (AGC) mechanism to adjust the gainresponsive to the received signal amplitude. The received signal is fedto AGC peak detector 36 which amplifies the signal and drives currentthrough diode 32 into capacitor 28 when the signal exceeds the AGCthreshold voltage V_(AGC) in order to generate a gain control signal.The gain control signal increases in response to increasing signalstrength and correspondingly reduces the gain of amplifier 26 so thatthe amplitude of the received signal at the output of amplifier 26remains relatively constant despite variations in received signalstrength.

At a minimum, infrared receiver 20 amplifies the photodetector signalcurrent and then level detects or demodulates the signal when it risesabove the detect threshold V_(DET) thereby producing a digital outputpulse at D_(OUT). For improved performance, the receiver may alsoperform the added functions of blocking or correcting DC and lowfrequency AC ambient (1-300 uA) signals and Automatic Gain Control (AGC)which improves both noise immunity and minimizes output pulse widthvariation with signal strength.

The structure of the conventional discrete PIN photodiode 24 isillustrated in FIG. 1B. A wafer 50 is lightly doped with N dopant inorder to produce an intrinsic region 56. A P+ region 52 is formed on onesurface of the wafer and an N+ region 58 is formed on the opposingsurface of wafer 50 with intrinsic region 56 interposed P+ region 52 andN+ region 58. A reflective layer 60, typically gold, is disposed on thesurface containing P+ region 58 with reflective layer 60 also serving asthe electrical contact to N+ region 58. A metal contact 54 is disposedon the surface containing P+ region 52 to provide the electricalconnection to the P+ region.

Typically, one power supply potential is applied to the reflective layer60 and another power supply voltage is applied to contact 54 to reversebias the PN junction formed by P+ region 52 and N+ region 18. This formsa large depletion region within the intrinsic region 56 wherein electronand hole charge carrier pairs generated by light photons incident uponthe intrinsic region 56 are rapidly accelerated toward the P+ and N+regions respectively by the electric field of the reverse bias voltage.Charge carrier pairs are also typically generated outside the depletionregion within intrinsic region 56 which diffuse, due to random thermalmotion of the carriers, at a much slower velocity until they reacheither the depletion region or the junction formed by P+ region 52 andintrinsic region 56 of photodiode 24.

A conventional photodiode that is designed for high quantum, i.e. lightconversion, efficiency requires that the light path within the photocurrent collection zone, i.e. the depletion and non-depletion zoneswithin intrinsic region 56, be sufficient in length so that most of thelight photons of the incident light signal area are absorbed andconverted into electron-hole pairs that are collectable at the P+ and N+regions. Usually, this requires that the width of the intrinsic region56, which is the primary light collection region, be several times thelength required for light absorption. If diode 10 has an efficientback-side reflector, such as reflective layer 60, which effectivelydoubles the light path within diode 24, then the intrinsic region 56 ofthe photodiode can be made narrower. For a typical near infrared siliconphotodiode, the nominal absorption path length is about 15-25 microns.The path length should be at least two to three times the nominalabsorption path length to obtain good light conversion efficiency.

On the other hand, a photodiode designed for high frequency responserequires that the photo current pairs generated by the light signal becollected rapidly and that the diode RC time constant is fast. Rapidphoto current pair collection usually requires that most of the photocurrent pairs generated by the light signal be generated with thedepletion region formed by the reverse bias voltage because the pairswill have a high drift velocity. Otherwise, the photo generated chargecarrier pairs produced in the non-depletion regions within intrinsicregion 56 and within diffusion distance of the collection electrodes 52and 58 will have a diffusion velocity that is several hundred timesslower than the velocity of the pairs generated within the depletionzone. The photo generated charge carrier pairs in the non-depletionzones will slowly migrate for collection at P+ region 52 and N+ region58 resulting in a tall on the trailing edge of the electrical signalcorresponding to the light signal. The diffusion distance of the chargecarriers is determined by the carrier mean free path beforere-combination and may exceed 150 microns.

A fast RC time constant for photodiode 24 requires minimal capacitanceand low series resistance between the electrical contacts 54 and 60 andthe photo current pair collection sites at the margin between P+ region52 and the depletion zone and the margin between N+ region 58 and thedepletion zone. The greater the width of the intrinsic region 56, thegreater the width of the depletion zone and the lower the capacitanceper unit area of photodiode 24. Since the width of the depletion zoneincreases with the magnitude of the reverse bias voltage, it is typicalfor high speed photodiodes to have a relatively high reverse voltageapplied to them.

The inclusion of lightly doped intrinsic region 56 between the P+ and N+regions 52 and 58 results in a PIN photodiode with a wider depletionregion, depending on the magnitude of the reverse bias voltage, whichimproves the light collection efficiency, increases speed, and reducescapacitance over that of a simple PN diode structure.

The PIN photodiode is typically produced by diffusing the N+ region 58on the back side of the lightly doped (N) wafer 50, diffusing the P+region 52 on the topside of the wafer 50, and then adding metal contactsto each side of the wafer. Typically, the backside contact areaconnected to N+ region 58 is reflective layer 60 and is made of gold.The reflective layer is then typically connected to the ground voltageterminal.

A PN diode junction can also function as a photodiode. However, thephoto-current collection region within an electric field, the driftregion, in a PN photodiode is limited to the relatively thin depletionzone produced when the PN junction is reverse biased. This thin driftregion is much less efficient in the collection of photo-generatedcharge carrier pairs because most of the pairs are generated outside ofthe depletion zone. Also, the charge pairs generated outside of thedepletion zone thermally diffuse to collection points margins of the Pand N layers and into the depletion zone at a much slower relative speedresulting in slow photodiode performance. In addition, the highly dopedP and N regions result in high diode capacitance per unit area whichfurther slows the performance of the photodiode.

Although a PIN photodiode outperforms a standard PN diode, the PINphotodiode structure cannot be easily manufactured by standardsemiconductor processes wherein fabrication is typically performed ononly one side of the semiconductor wafer 50.

In typical high volume applications, it is now standard practice tofabricate the receiver circuitry and transmitter driver in a singleintegrated circuit (IC) to produce a transceiver IC. As described above,it is difficult to integrate an efficient photodiode on the samesemiconductor substrate as the transceiver circuit. As a result, adiscrete infrared photodiode is typically assembled with the transceivercircuit and an LED, along with lenses for the photodiode and LED, into aplastic molded package to form a transceiver module. The transceivermodule is designed to be small in size and allow placement in theincorporating electronic device so as to have a wide angle of view,typically through an infrared window on the transceiver casing. Thetransceiver IC is designed to digitally interface to some type of serialdata communications device such as an Infrared Communication Controller(ICC), UART, USART, or a microprocessor performing the same function.

SUMMARY OF THE INVENTION

The present invention relates to a distributed photodiode which can befabricated using standard semiconductor fabrication processes and whichincorporates dopant gradients that effect the migration of chargecarriers.

An embodiment of a distributed photodiode, according to the presentinvention, is composed of a substrate doped with a first dopant type andhaving first and second planar surfaces. The photodiode includes a firstplurality of diffusions doped with a second dopant type and formed onthe first planar surface of the substrate. A second plurality ofdiffusions of the photodiode are doped with the first dopant type andformed on the first planar surface of the substrate interposed the firstplurality of diffusions. A first contact having a first plurality ofconnective traces is disposed on the first planar surface of thesubstrate and coupled to each of the first plurality of diffusions.

An embodiment of a method for producing a photodiode, according to thepresent invention, involves providing a substrate having first andsecond planar surfaces, doping the substrate with a first dopant type,and diffusing a second dopant type at a plurality of sites on the firstplanar surface of the substrate to form a first plurality of diffusions.The method further recites diffusing the first dopant type at anotherplurality of sites on the first planar surface of the substrate to forma second plurality of diffusions. The method also calls for disposing afirst set of conductive traces on the first surface planar surface tointerconnect each of the first plurality of diffusions.

Another embodiment of a photodiode, according to the present invention,includes a substrate doped with a first dopant type and having first andsecond planar surfaces and a first plurality of diffusions, the firstplurality of diffusions being doped with a second dopant type and formedon the first planar surface of the substrate. The photodiode alsoincludes a backside diffusion formed on the second planar surface of thesubstrate, the backside diffusion being doped with the first dopanttype. The photodiode also includes a first contact having a firstplurality of connective traces disposed on the first planar surface ofthe substrate and coupled to each of the first plurality of diffusions.

The features and advantages of the present invention will become morereadily apparent from the following detailed description of a preferredembodiment of the invention which proceeds with reference to theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following drawings, elements which are identical or analogousbetween drawings are identified with reference numbers which are alsoidentical or analogous.

FIG. 1 is a circuit diagram of a conventional infrared transmitterreceiver pair;

FIG. 2 is a cross-sectional diagram of a conventional PIN photodiode;

FIG. 3 is a diagram of an embodiment of an integrated receiver and aphotodiode according to the present invention;

FIG. 4A is a cross-sectional diagram of a portion of the photodiode ofFIG. 3;

FIG. 4B is a top-view diagram of a portion of the photodiode of FIG. 3;

FIG. 5 is a diagram of a n embodiment of the integrated receiver andphotodiode of the present invention wherein the photodiode includes aconductive shield;

FIG. 6 is a functional block diagram of one embodiment of an infraredreceiver circuit suitable for use in the integrated receiver andphotodiode of the present invention;

FIG. 7 is a functional block diagram of another embodiment of aninfrared receiver circuit suitable for use in the integrated receiverand photodiode of the present invention;

FIG. 8 is a functional block diagram of yet another embodiment of aninfrared receiver circuit suitable-for use in the integrated receiverand photodiode of the present invention;

FIG. 9 is a functional block diagram of still yet another embodiment ofan infrared receiver circuit suitable for use in the integrated receiverand photodiode of the present invention;

FIG. 10 is a functional block diagram of a further embodiment of aninfrared receiver circuit suitable for use in the integrated receiverand photodiode of the present invention;

FIG. 11 is a diagram of an embodiment of the integrated receiver andphotodiode of the present invention utilizing a photodiode having anadditional set of insulated traces for use with a differential inputreceiver;

FIG. 12 is a cross-sectional diagram of an embodiment of a photodiode,according to another aspect of the present invention, having surfacediffusions to form a dopant gradient;

FIG. 13 is a cross-sectional diagram of an embodiment of a photodiode,according to another aspect of the present invention, including abackside diffusion to form a dopant gradient; and

FIG. 14 is a cross-sectional diagram of an embodiment of a photodiode,according to another aspect of the present invention, where the surfacediffusions are formed in trenches.

DETAILED DESCRIPTION OF THE PRESENT INVENTION

Because of the cost associated with assembling the transceiver moduleinto a single package, it is desirable to integrate the photodiode andthe receiver circuit on a single silicon substrate. However,constructing an IrDA receiver or transceiver with an integratedphotodiode has typically not been either cost effective or sufficientlyoptically sensitive to have the range to meet, for example, the IrDAlowspeed specification.

In general, IrDA receivers typically use discrete photodiodes with areasof 3.4 to 25 square millimeters in order to produce the minimum signallevel, typically 30 nA to 220 nA, required by the minimum detectthreshold of an IrDA receiver at a range of 1 meter. A receiver ortransceiver IC typically has an area of 2 to 10 square millimetersdepending upon the IC fabrication process technology employed and thecomplexity of the circuit design.

Although PN photodiodes can be constructed using a standard ICfabrication process, the resulting photodiodes typically suffer fromreduced efficiency, signal bandwidth, or very high capacitance ascompared to PIN photodiodes fabricated using a process optimized forphotodiodes. Because there are significantly fewer processing steps andfewer photo-lithographic masks of low optical resolution required for aphotodiode optimized process, a PIN photodiode specific silicon processtypically costs about one half or less per unit area than a modemsilicon IC process. In addition, because process yields are higher andtesting is simpler for a photodiode specific process, there is a furtherreduction in cost per area of perhaps another 20% to 40%. However, theprocessing and yield cost advantages of the photodiode specific processare partially offset by the assembly cost incurred in the handling,inventory, die attachment, and bonding of the discrete photodiode into areceiver or transceiver module. Also, using conventional design andfabrication techniques, the photodiodes in an IrDA receiver ortransceiver are significantly larger than the receiver or transceiver ICand the photodiodes can be fabricated as discrete devices at lower costper unit area. Thus, it is more cost effective not to integrate thephotodiodes with the receiver or transceiver, despite the savingsobtained by eliminating the photodiode assembly cost.

Furthermore, PIN photodiodes constructed using a standard IC processwill have a light conversion efficiency per unit area which is typically20% to 80% of the efficiency of a PN photodiode fabricated using aphotodiode specific process. Utilizing the assumption that thephotodiode processing costs are typically one half that of the ICprocess, the integrated diode must be less than half the area of anexternal photodiode in order for an integrated photodiode and receiveror transceiver IC to be more cost effective than a discrete photodiodesolution. The integrated photodiode will therefore have a signal outputthat is only 10% to 40% of the signal output of a cost-equivalentdiscrete PIN photodiode.

There are still other problems with photodiodes fabricated usingstandard IC processes. One major problem is that a photodiode fabricatedon a standard IC process typically has a capacitance per unit area thatis over 100 times higher than the capacitance per unit area of aphotodiode fabricated using a process optimized for photodiodes. Thisincreased capacitance causes a significant reduction in the infraredreceiver noise/bandwidth performance which in turn requires the use of alarger integrated photodiode than would otherwise be required.

The smallest photodiode signal current which can be detected, given aparticular receiver bandwidth, must be greater than the equivalent inputnoise level of the receiver. Hence, if the receiver's noise level can bereduced then a smaller received signal can be detected. The lower boundon an IC receiver's equivalent input noise level is limited by the inputtransistor noise performance and area, supply voltage, input device biascurrents and the source impedance, which is determined by signalfrequency and photodiode capacitance. The principles of low noisephotodiode amplifier design, which govern the input noise parameters,are well known to those skilled in the art and will not be expanded uponhere. However, given a particular bandwidth, it should be understoodthat as the capacitance of the photodiode increases and its impedancedecreases, then the equivalent input noise current will increase as asquare root function, assuming that the infrared receiver amplifier isoptimized for minimum noise under these conditions.

For example, a four times increase in the input capacitance of thephotodiode will typically result in a two times increase in theequivalent input noise current of a receiver including the photodiodewhen the receiver is optimized for minimum noise within the samebandwidth. Notice, however, that increasing the area of a photodiode byfour times increases the output current gain by four times although itonly increases the input noise by two times. Consequently, increasing aphotodiode area increases its effective gain, or signal-to-noise ratio,for a given signal illumination level. However, it is also clear that,for a photodiode which has a capacitance that is 100 times greater perunit area, the receiver input noise current of an optimized receiver isincreased by 10 fold and therefore requires the minimum detect thresholdof the receiver to be increased by 10 fold. Thus, although a photodiodecan be constructed using a standard IC process to have only a moderatelydegraded efficiency as compared to a photodiode constructed using aphotodiode optimized process, the 100 fold larger capacitance of thestandard IC process photodiode degrades the receiver noise performanceso as to effectively reduce input sensitivity by 10 fold.

Another problem is that an efficient photodiode typically uses thesubstrate as one of the diode contacts because the light absorptiondepth of the silicon typically exceeds by several times the shallowjunctions on standard IC processes. Since the substrate contact istypically P material, this means that the substrate is an anodeconnection, which is the reverse of standard external photodiodestructure which usually have cathode connections to the substrate.Consequently, an integrated photodiode receiver typically needs to beable to function with the anode to substrate, which is usually coupledto a ground potential supply rail.

In view of these constraints, it is nonetheless possible to build anIrDA receiver with sufficient sensitivity and bandwidth to operate usinga small photodiode integrated on the same IC. As an example, such areceiver might use a cost effective integrated photodiode of about 2 mm²in area. Assuming that the integrated photodiode has about 60% of theefficiency of a standard discrete photodiode, the integrated photodiodewould output about a 18 nA pulse when suitably lensed and illuminatedwith 4 uW/cm² of infrared light, which is the minimum infraredirradiation level specified for IrDA slow speed operation with a 10⁻⁸bit error rate. The receiver utilizing the integrated photodioderequires an equivalent input noise level over the pulse bandwidth ofabout 2 nA and a signal detection threshold of about 9 nA.

A major problem for the design of IrDA infrared receivers is outputfeedback noise. The integrated photodiode is vulnerable to feedbackcapacitance from the receiver output D_(OUT) to the photodiode. Forexample, an integrated photodiode of 2 mm² will have a feedback couplingcapacitance in the range of 2 to 10 femto Farads (fF) between the uppersurface of the photodiode and the surface of the receiver output bondingwire, which will typically be less than 2 mm distance away. Although theintegrated photodiode is smaller than a discrete external photodiode,the integrated photodiode is necessarily located closer to the receiveroutput because it is on the same IC as the receiver circuit.Consequently, the feedback coupling capacitance is of the same order ofmagnitude as an external photodiode. In addition, because the receivermust be more sensitive to compensate for the smaller and less efficientintegrated photodiode, the effects of feedback are proportionatelylarger.

Specifically, taking the above examples, if the feedback capacitance isas little as 2 femto Farad and the receiver output has a 5V swing, then10 femto Coulombs (fC) of charge will be transferred to the receiverinput terminal for each transition at the receiver output terminal. Ifthe receiver bandwidth is 600 kHz, which is a typical value for a 115Kbps lowspeed IrDA receiver, then the time constant of the signaltransient is ({fraction (1/600)} kHz)/6.28 or about 265 nsec.Consequently, the 10 fC of charge will have an apparent currentamplitude of (10 fC/265 nsec) or 376 nA. Notice that this is over 40times the minimum detect threshold of 9 nA in the above example and willcause disruption of the input signal over 80% of the operating range ofthe receiver.

Some of the conventional remedies for output to input feedback problemsof IrDA, receivers are: 1) to use a large photodiode so that the outputfeedback is proportionately small compared with the signal; 2) installshielding between the output and the input photodiode; and 3) balanceddifferential output lines and/or balanced input photodiodes. All ofthese solutions entail increased cost either due to increases in totalphotodiode area, increased package complexity, or the cost of adding ashield.

In addition, for an IrDA receiver or transceiver IC incorporating anintegrated photodiode, adding a shield between the photodiode and thereceiver output is especially difficult. Since the IC is very small,typically about 3 to 4 mm long, the shield needs to be placed accuratelywithin 1 mm after die attach and wire bonding. This extra assemblyprocess adds cost to the final package. Another difficulty is that theshield must be designed not to block the infrared light falling on theintegrated photodiode.

Another receiver output to photodiode input feedback mechanism notpresent with an external photodiode but which may be present with aninfrared receiver integrated with an on-chip photodiode is feedbackcoupling through the substrate. This may be due to either carriersinjected by the receiver circuitry or by distributed RC coupling throughthe substrate. For this feedback coupling mechanism, external shieldingwill not work, although some benefit can be had by placing diffused Pand N collector rings connected to suitable supply voltages. These acteither as limited substrate shields or as collectors for substratecarriers.

Finally, another problem for infrared data receiver performance, whetherbuilt with an external or an integrated photodiode, is external noisepickup, most notably from adjacent digital signal lines. Although themagnitude of the external noise signal is typically less than themagnitude of the feedback signal from the receiver's own outputterminal, the external noise can cause receiver disruption from digitallines or other close-by circuit nodes with significant AC signal levels.External shields are often used to control noise from external signals,which again increases the cost of the package and/or increases its totalsize.

FIG. 3 illustrates an embodiment of an integrated receiver andphotodiode 300 according to the present invention. To reduce thecapacitance per unit area arising from the shallow junctions of standardID processes, a photodiode 330 is constructed by forming the PNjunctions of photodiode 330 from N-diffusion dots 332 spaced from oneanother and connected to one another using conductive traces 334.(N-diffusion dots 332A, 332B and 332C will be discussed in detail butare illustrative of the totality of N-diffusion dots 332 which make upphotodiode 330.) The connective traces 334 are coupled to the inputterminal D_(IR) of receiver 220. Again, as with photodiode 230, theoutput terminal D_(OUT) of receiver 220 is capacitively coupled,represented by capacitor 40, to photodiode 330 and a feedback controlreceiver design can be utilized to reduce the effects of feedback. Inaddition, there is capacitive coupling to external noise sources, whichis represented by capacitor 42.

FIG. 4A is a partial cross sectional view of an embodiment of thephotodiode 330 of FIG. 3. Multiple N-diffusions 332A, 332B and 332C areformed in P-substrate 210 at regular intervals to form the PN junctionsof photodiode 330. The N-diffusions are formed from minimum geometrydots of width W spaced at a uniform distance D from each other and arethen connected together with minimum capacitance interconnects 334A,334B and 334C, respectively, which can be formed using upper metal layerminimum geometry traces. The junction dot spacing D is determined by themean free carrier diffusion path and the maximum pulse decay timeconstant for the receiver. For example, if the mean free carrierdiffusion path length is 155 microns and the diffusion velocity ofcarriers is 300 m/sec and if the N-diffusion dots 332A-C are separatedby 50 microns, then the maximum diffusion distance of light generatedcarriers to a junction dot is about 25 microns, i.e. 25 microns dividedin half. This means that the pulse decay time constant is approximately1.2 usec, the time for the carriers to diffuse 25 microns.

In a typical 0.8 micron BiCMOS process, the minimum diffusion dot sizeis approximately 2×2 microns. If N-diffusion dots 332A-C are placedevery 50×50 microns, then the diffusion junction of photodiode 330occupies less than {fraction (1/500)} of the area of the photodiode. Theresulting capacitance from the N-diffusion dots 332 to substrate 210 ofphotodiode 330 is about {fraction (1/80)} of a comparably sizedphotodiode having a continuous junction. The capacitance to substrate210 is larger than the junction area ratio due to the added capacitanceof the N-diffusion sidewalks and the capacitance of the minimum geometrymetal interconnects 334A-C.

FIG. 4B is a top view of photodiode 330 which illustrates thedistribution of N-diffusion dots on a surface of P-substrate 210.N-diffusion dots 332A-C are spaced at uniform distance D from oneanother. All the N-diffusion dots are equally spaced and connected viainterconnects 334A-C to form photodiode 330.

The distributed structure of photodiode 330 tends to extend the durationof the infrared pulse by about one half the carrier decay time constant,thus it is beneficial to reduce the output pulse width by this amount torestore the output pulses to the input pulse width. The spacing D of theN-diffusion dots 332 can be varied in order to adjust the response speedof the photodiode 330. In addition, commonly assigned, co-pending U.S.patent application Ser. No. 08/864,286 entitled APPARATUS AND METHOD FOROUTPUT SIGNAL PULSE WIDTH ERROR CORRECTION IN A COMMUNICATIONS RECEIVER,now U.S. Pat. No. 6,169,765, herein incorporated for all purposes,describes a circuit and method which can be employed to normalize thepulse-width of the output pulse of receiver 220.

The connective traces 334 shown in FIGS. 3 and 4A are typically metalstrips that are approximately 1 micron wide and spaced at 43 micronintervals. The metal of the connective traces 334 occupies approximatelyV40 of the amount of surface that would be covered by a conductive layerfor a conventional continuous diode. As a result, the connective traces334 present a smaller antenna for noise signal reception and thecapacitive coupling to the feedback signal from the output terminalD_(OUT) and external noise sources is also reduced.

An advantage of the N-diffusion dot method illustrated in the photodiode330 of FIGS. 4A and 4B is that its effective coupling capacitance toexternal noise sources and internal substrate RC coupling effects issignificantly reduced. (However, it does not provide immunity fromsubstrate carrier noise injected by circuits of the receiver ortransmitter also' fabricated on substrate 210.) In photodiode 330, thecoupling capacitance is on the order of {fraction (1/50)} of the area ofthe photodiode which results in a significant improvement in the signalto noise ratio of the integrated receiver.

The exposed portions of substrate 210 between the N-diffusion dotconnections 334A-C also act as an effective shield with the addition ofgrounded backplane 212 to photodiode 330 shown in FIG. 4A. Without thebackplane, the screen created by the metal interconnect would havesimilar coupling capacitance to external noise sources as a continuousconductor over the same area. However, in the present invention, thegrounded backplane 212 sets up an inverted induced noise signal whichneutralizes the noise signal in the field around connective traces 334.As a result, the only noise signal that remains to generate a noisedifference signal in a receiver circuit is the {fraction (1/40)}th ofthe area of the backplane 212 that is shadowed by the connective traces334.

Although the total area of the N-diffusion dots 332 of photodiode 330 isabout {fraction (1/40)}th the area for a comparable photodiode having acontinuous N-diffusion region, the efficiency of the N-diffusion dots332 in absorbing thermally migrating photo-generated charge carrierpairs within P-substrate 210 is only about 20% less than that of acontinuous N-diffusion region. The heavily doped PN junction formed indiodes formed using standard semiconductor processes result in adepletion layer at the PN junction of only 1 micron. The absorptiondepth of the incident light, however, is approximately 15 microns. As aresult, there are a large number of photo-generated charge carrier pairsproduced in P-substrate 210. However, the grounded P-substrate 210effectively shields these carriers from electrical fields and so thecarriers thermally wander, i.e. the carriers are subject to Brownianmotion, without the effect of electrical fields upon them.

Thus, each of the photo-generated carriers in P-substrate 210 can beviewed as being at the center of a spherical cloud representing theprobability distribution for the thermal migration of that carrier. Thisspherical cloud has a radius of approximately 150 microns, the averagedistance travelled by the carriers before recombination. Thus, eventhough the carriers typically travel a longer distance to reach a PNjunction formed by one of the N-diffusion dots 332 than the distancethat would be travelled in a photodiode with a continuous PN junction,the probability that each carrier will reach the PN junction of one ofthe N-diffusion dots 332 before recombining is only slightly reduced.Thus, the distributed structure of photodiode 330 is only marginallyless efficient that a comparable photodiode having a continuous PNjunction while the capacitance of photodiode 330 is significantlyreduced.

FIG. 5 illustrates the integrated photodiode 330 and receiver 220 shownin FIG. 3 with the addition of a conductive shield 502. The shielding ofphotodiode 330 can be improved with only a modest increase in photodiodecapacitance by disposing conductive shield traces 502A-D above theN-diffusion dots 332 and interconnect traces 334. The conductive shieldtraces 502A-D are then coupled to a ground potential to thereby groundany noise signal that is incident upon the conductive shield 502. Theconductive shield traces 502A-D are typically fabricated using a secondmetal layer of the semiconductor fabrication process. The addition ofconductive shield traces 502A-D to photodiode 330 increases thephotodiode capacitance by approximately 50%. However, conductive shield502 also shields approximately 10 dB of the external noise signalincident upon photodiode 330.

The efficiency and low coupling capacitance of photodiode 330 may behigh enough to permit the use of a conventional receiver circuit such asreceiver 20 of FIG. 1 as receiver 220 fabricated on the same substrate.In that case, photodiode 330 would be substituted for photodiode 24 inthe circuit of FIG. 1, but both photodiode 330 and receiver circuit 20would be fabricated on the same semiconductor substrate.

However, the close proximity of the data input D_(IR) and D_(OUT) of areceiver and photodiode pair fabricated on the same substrate, with theeffect of additional coupling taking place through the substrate, maymake a feedback mitigation receiver circuit design desirable for copingwith the feedback signal from D_(OUT) to D_(IR). FIG. 6 illustrates oneembodiment of a receiver circuit 420 designed to mitigate the effects offeedback through control of the feedback phase with the design of thereceiver bandwidth filter design such that receiver circuit 420 issuitable for use as the receiver circuit 220 of FIG. 3. For infraredreceivers which demodulate using on-off modulation (such as themodulation specified by IrDA), it is possible to receive signalssignificantly below the feedback transient amplitude provided that thefeedback is in phase with the received signal. This is accomplished bydesigning the receiver so that the feedback from the data output ispositive such that the feedback actually reinforces the received signalbecause the polarity of the feedback spike corresponds to the polarityof the received signal. If the receiver transient response has littleovershoot, and either no AGC or high signal threshold AGC is used, thenthe positive feedback acts as dynamic hysteresis, producing an outputpulse without any spurious transitions.

The infrared receiver 420 of FIG. 6 is designed so that the feedbackthrough parasitic capacitor 440 from D_(OUT) to the photodiode inputD_(IR) is positive. Thus, a negative transition in the received signalat D_(IR) results in a negative transition in the data output signal atD_(OUT) which will, in turn, generate a corresponding negative spike inthe feedback signal.

The received signal is countered by the much larger amplitude of thefeedback spike which causes the signal at the input of detect comparator442 to cross the detect threshold V_(DET) repeatedly. This results inmultiple transitions in data output signal thereby corrupting the data.When the receiver is designed for positive feedback, as is receiver 420,the positive feedback reinforces the received signal. The positivefeedback, when combined with the received signal at D_(IR), results in asignal at the input of comparator 442 that swings farther away fromV_(DET) responsive to the edges in D_(IR) generating a single outputpulse at D_(OUT).

Bandpass filter 434 must be designed for good damped transient responseto suppress signal ringing and overshoot. An example of a suitablefilter is a Gaussian bandpass filter, which has roll-off edges in thesignal output from the filter that have low ringing and overshoottransient response. In addition, the filter will temporally spread outthe energy contained in the feedback spike. For example, a 20 nsec.spike is transformed into a 300 nsec. spike, which further contributesto the dynamic hysteresis discussed above.

However, in order to prevent AGC desensitization, the AGC thresholdV_(CC) of receiver 420 needs to be set above the peak feedback value bya margin adequate to prevent the peak feedback value from causing thegain to be adjusted downward by the AGC. There are undesirableconsequences of a high AGC threshold compared with a low AGC threshold.First, AGC noise quieting (which reduces signal interference from noise)is less effective. Secondly, the output pulse width will vary more withsignal level.

AGC reduces the front end gain of amplifier 426 responsive to anincreasing input signal on D_(IR). Generally, in the absence of an inputsignal, amplifier 426 will be highly sensitive because the AGC permitsthe gain to be high. In the presence of noise, however, the AGC willreduce the gain in response to the input signal including the noiselevel. This improves the noise immunity of the receiver 420. Thereceiver will function so long as the received signal strength isgreater than the amplitude of the noise. From a practical standpoint, ina noisy environment, this permits the sending and receiving devices tobe moved closer together to strengthen the received signal and thecommunications link will be able to function. In the absence of AGC, thenoise level will prevent the receiver from capturing the transmittedsignal without corruption of the output data signal even when thetransmitting device and receiver are close together because the noisesignal will still have high enough amplitude to cause spurious outputtransitions in between valid output transitions.

Also, AGC improves the fidelity of the pulses in the data outputD_(OUT).

Despite careful filter design, some ringing, overshoot and undershootwill still occur in the receiver. AGC reduces the effect of ringing,overshoot and undershoot when it reduces the sensitivity of amplifier426. Further, because there is also ramping on the received waveformwhich can cause widening or narrowing of the signal pulse unless thedetect threshold V_(DET) is in the center of the waveform, AGC improvesthe fidelity of the pulse by maintaining V_(DET) at the center of thewaveform.

The amplitude of the positive feedback, however, also causes the AGC toadjust the receiver sensitivity downward. As the AGC reduces the gain inresponse to the positive feedback, the sensitivity of the receiver tothe transmitted signal is also reduced and, particularly at high signalpulse rates, can cause the receiver to lose the input signal.

Whereas the receiver circuit of FIG. 6 is effective in reducing theeffects of feedback, it still suffers from the effect of feedbacktransient overshoot or ringing, which, if it exceeds the detect levelV_(DET), will cause undesirable extraneous output pulse transitions.Although the use of well known filter design techniques can limittransient overshoot to a negligible level, in practice, reducing it to avalue below ⅕ or {fraction (1/10)} the peak level is difficult due tovariable phase shift effects both within and outside the infraredreceiver. Some of these variable phase shift effects are due to normalvariances in parameters such as transmit pulse shape, photodiode timeconstant, photodiode capacitance, receiver output load capacitance,receiver supply voltage, and filter component values.

Receiver 420 can beneficially decrease the disruptive effects offeedback by 10 db-20 db for infrared receivers used with edge triggered,serial data communication controllers which do not need an accurate datapulse width or with receiver systems which do not require the benefitsof a low threshold AGC.

Another embodiment of a receiver suitable for use in the presentinvention is shown in FIG. 7. Infrared receiver 620, which is suitablefor the use of low threshold AGC, adds delay 650 to the output signalfrom comparator 642, typically delaying the output by ½ of the pulseinterval of the data signal to permit decay of the feedback pulse.Receiver 620 also includes a signal disable switch 648 controlled by AGCdisable one-shot 652 which blocks the signal to the AGC input upon theleading edge of an output pulse transition on D_(OUT).

By delaying the output signal, the peaks of the feedback signal fromD_(OUT) to D_(IR) are shifted in time so that the feedback peak occurstoward the center of the pulse in the received signal at D_(IR). Also,AGC disable one-shot 652 generates a signal disable pulse responsive tothe falling edge in the output signal at D_(OUT) which causes AGCdisable switch 648 to open and isolate the input of AGC peak detector636 from the received signal path for the duration of the signal disablepulse. The signal disable pulse must persist for a time intervalsufficient for the feedback transient to settle below levels which wouldcause AGC gain reduction. AGC peak detector 636 is therefore isolatedfrom the signal path at the time that the feedback pulse appears at thenegative input to comparator 642. As a result, the gain control voltagestored in capacitor 628 during the disable period reflects the receivedsignal strength and is not corrupted by the feedback signal.

Alternatively, the AGC disable one-shot 652 may be replaced with anoutput edge triggered disable one shot which will generate a disablepulse responsive to both the falling and rising edges of the outputsignal at D_(OUT). An edge triggered disable will isolate the AGC peakdetector 636 during feedback pulses for both the edges of the outputpulse at D_(OUT). This approach has the advantage that larger levels offeedback can be tolerated or the use of a low AGC threshold voltagelevel is permissible because the gain control voltage is not adverselyaffected by the feedback pulse from the rising edge of the output signalat D_(OUT).

Whereas the performance of receiver 620 is substantially better than theperformance of conventional receivers, it requires that receiver 620 bedesigned so that the feedback from D_(OUT) to D_(IR) is positive.Negative feedback will still cause spurious transitions in the outputsignal because the feedback is coupled to the input, of comparator 642.

Receiver 820 of FIG. 8 shows yet another embodiment of a single inputreceiver circuit suitable for use as the receiver 220 in the presentinvention but where the design of receiver 820 is not dependent uponpositive feedback from output terminal D_(OUT) to input terminal D_(IR)signal disable switch 848 is positioned between the output of bandpassfilter 834 and the inputs of both comparator 842 and AGC peak detector836. Output edge triggered disable one-shot 852 receives the delayedoutput signal at D_(OUT) and generates a disable pulse responsive toeach of the falling and rising edges of the output signal. The disablesignal must persist for a period long enough for the feedback transientsto decay below a level that would cause spurious transitions in theoutput signal from comparator 842. The disable pulses cause signaldisable switch 848 to isolate comparator 842 and AGC peak detector 836from the received signal path during the times when the falling andrising feedback peaks are present at the output of bandpass filter 834.This configuration permits receiver 820 to obtain the improved AGCperformance of receiver 620. However, receiver 820 is not dependent uponpositive feedback because the input of comparator 842 is isolated fromthe feedback peaks, thereby preventing the negative feedback peaks fromcausing spurious transitions in the output signal at D_(OUT).

FIG. 8 also shows greater detail of an example of an output edgetriggered disable one-shot. One input terminal of exclusive-OR gate 852receives the output signal directly while the other input terminal iscoupled to the output signal through resistor 856 and to ground throughcapacitor 854. Resistor 856 and capacitor 854 further delay the outputsignal such that, when a pulse edge occurs in the output signal, the twoinputs of XOR 852 will be at different values for a time perioddetermined by the RC constant of resistor 856 and capacitor 854, whichtherefore also determine the width of the disable pulses generated byXOR 852.

Yet another embodiment of a receiver circuit suitable for use asreceiver 220 of FIG. 3 is receiver circuit 920 of FIG. 9. The highamplitude of the feedback pulses can contributes to ringing, overshootand undershoot at the output of bandpass filter 834 of receiver 820.Thus, receiver 920 is constructed with signal disable switch 948interposed between the output of input amplifier 926 and bandpass filter934. This configuration permits the disable pulses generated by XOR 952responsive to the edges in the output signal to isolate bandpass filter934, comparator 942 and AGC peak detector 936 from the receive signalpath when feedback from D_(OUT). This configuration causes the feedbacktransient to settle more rapidly because the transient is prevented fromentering the bandpass filter 934 where the transient is prolonged due tothe increase in ringing, overshoot and undershoot that would otherwiseoccur due to feedback transients in the output response of bandpassfilter 934. This permits receiver 920 to tolerate larger amplitudefeedback signals, use a narrower bandwidth to improve the signal tonoise ratio of the receiver, or operate at faster pulse rates withoutcorruption of the data output signal at D_(OUT).

Because of variations in device tolerances and operating conditions, itis not possible to exactly predict when the feedback transients in thereceived signal path will settle. For slower communications formats, theduration of the disable pulses may be extended to account for variationsin the settling time of the feedback peaks. However, fastercommunications formats, such as the 4 MB format described above, havenarrow windows because the data is related to the temporal position ofthe pulse which requires rapid settling times. As a result, the disablesignal generated by XOR 952 may not coincide exactly with the feedbackpeaks.

For example, if the disable signal is 1.5 us, then valid input signaltransitions which have less than 1.5 us between them cannot be detected.To capture these signal transitions, it is necessary to set the disablesignal duration to the minimum required to prevent feedback disruption.However, due to variances in IC timing circuit tolerances and variancesin feedback due to variances in receiver packages and circuit boardtrace layout, it becomes necessary to set the signal disable period to alarger value than is typically required so as to ensure that mostreceivers will function without feedback disruption. This adds adifficult engineering burden of correctly determining the optimum signaldisable duration and undesirable limiting maximum pulse rate on receiverpackages or board layouts which have low feedback levels.

Yet another embodiment of a receiver circuit suitable for use asintegrated receiver 220 is receiver 1020 of FIG. 10. In order toaccommodate faster communications formats and address variation insignal settling, receiver 1020 is designed with positive feedback andfeedback detection which monitors the feedback transients and preventssignal disable switch 1048 from closing while a feedback transient ispresent.

Feedback detect comparator 1060 compares the amplified input data signalat the output of input amplifier 1026 to the detection threshold voltagelevel VDET and output a high level signal so long as the amplified inputsignal is greater that the detection threshold. Exclusive-OR 1062compares the output of the feedback detect comparator 1060 with thedelayed digital output signal and outputs a high level signal if the twosignals differ. Conversely, exclusive-OR 1062 outputs a low level if thetwo signals are in agreement. This low output signal will propagatethrough AND gate 1064 to close signal disable switch 1048 before theedge triggered disable signal output from exclusive-OR 1052 wouldnormally cause switch 1048 to close. This permits receiver 1020 tooperate at higher speeds when the transient settling time is faster thanthat predicted solely by the timing of the edge-triggered one shotcircuit.

The photodiode of the present invention can also be incorporated intodifferential methods for receiving an infrared signal. FIG. 11 showsanother embodiment of a photodiode 1230 according to the presentinvention which can be employed with a receiver 1120 having a pair ofdifferential inputs D_(IR), and D_(IR2). Photodiode 1230 is similar instructure to photodiode 330 of FIG. 3. As with photodiode 330, a seriesof N-diffusions 332 are formed in P-substrate 1210 and interconnectedvia connection traces 1234 to input terminal DIR, of receiver 1120. Anadditional set of insulated traces 1238 is formed adjacent theconnection traces 1234 and connected to input terminal D_(IR2) ofreceiver 1120. Because connection traces 1234 and insulated traces 1238have the same area, they will each receive the same noise signal whichis subsequently cancelled by the differential nature of the receiver1120. Since the insulated traces 1238 occupy only about V50 of the areaof the photodiode, they only slightly reduce the light efficiency of thephotodiode 330. Because the insulated traces have the same area asconnection traces 1234, they will pick up approximately the samemagnitude of noise signal. To maximize the differential balance betweenthe connection traces 1234 and the insulated traces 1238, the insulatedtraces should have the same structure and the connection traces andshould be interleaved with the connection traces 1234 as is shown inFIG. 11.

The receiver circuits 420, 620, 820, 920 and 1020 can each be modifiedto operate in a differential mode with the photodiodes illustrated inFIG. 11, as should be apparent to one skilled in the art.

The PN dotted photodiode disclosed in U.S. patent application Ser. No.09/037,258, now U.S. Pat. No. 6,198,118 B1, herein incorporated byreference in its entirety for all purposes, teaches how to produce aphotodiode having high electric field immunity, low manufacturing costs,and that allows integration of the photodiode on the same integratedcircuit (IC) fabrication process used for manufacturing the transceivercircuit. Although this photodiode has sufficient speed performance foruse in infrared communication systems that operate at up to 1-2megahertz of bandwidth, it may not be fast enough for some higher speedapplications or systems. For these systems, a conventional PINphotodiode is typically required for adequate speed performance. Suchhigher speed PIN photodiode systems also require either specialshielding against interference from ambient noise or differential diodetechniques that generally use a larger amount of silicon area. It isdesirable to improve the speed performance of the PN dotted photodiodestructure without sacrificing its high electric field immunity or lowcost of manufacturing.

For photodiodes that are intended for data communication, the keyperformance parameters are bandwidth, optical signal gain per unit area,and low capacitance. Bandwidth typically limits the maximum speed ofdata transmission, optical signal gain typically affects sensitivity,and low capacitance is important to minimize noise. See the discussionabove for further explanation of some of these issues.

As discussed above, the bandwidth of the PN dotted photodiode is limitedby the diffusion speed of minority carriers to the diffusion dots.Reducing the diffusion dot distance increases speed or bandwidth forminority carriers generated by photon absorption near the surface butdoes not help much for those carriers generated at silicon depths morethan about one-half the dot spacing. Dot spacing less than two times theoptical absorption depth generally creates three performance problems.First, the high bandwidth optical gain is typically reduced. Secondly,the total capacitance of the photodiode is increased due to theincreased number of diffusion dots and inter-connect metal. Finally, theoptical gain is further decreased due to light shielding by opaqueinterconnect metal.

One partial remedy for the problems involving optical gain andcapacitance caused by reducing the dot spacing is to fabricate thephotodiode diffusion dots and interconnect metal with smallerdimensioned lithography. Although this can reduce capacitance, fringingeffects limit the capacitance reduction obtained with diminishing linewidths for line widths of less than the field oxide thickness. Forexample, typical field oxide thickness is about 1 micron. This meansthat once a line width falls below 1 micron, then there is littlereduction in line capacitance for thinner lines. Increasing oxidethickness would reduce capacitance but field oxides thicker than 1micron are problematic in that they are more expensive to produce andcause yield problems from the large metal steps required to connectcontacts on the surface of the field oxide to the active regions of thephotodiode. Similarly, a reduction in line width reduces light shieldingeffects, but, due to diffraction effects, these improvements are alsolimited.

Another problem is that the silicon foundries which are most costeffective for manufacturing the PN dotted photodiode are typically notcapable of lithography line widths of less than several microns. Siliconfoundries suitable for photodiode production tend to specialize inproducing silicon products that have low costs per square millimeter;such as, high voltage devices or solar cells, neither of which requiresub-micron lithography. Most sub-micron lithography silicon foundriesuse 20 or more masks to produce very complex products with highfunctionality or value per square millimeter to offset thecorrespondingly high costs per square millimeter. Consequently, althoughit is possible to fabricate the PN dotted photodiode with sub-microndiffusion dots and metal inter-connect traces, the costs would tend tobe higher and may require unconventional process changes that are notreadily accommodated by most modem silicon foundries.

In the PN dotted photodiode according to the present invention, absorbedphotons generate electron hole carrier pairs. The minority carrierportion of these pairs travels by diffusion transport to the diffuseddots that are of opposite polarity doping to that of the substrate wherethey are collected, which causes current to flow that is proportional tothe photo flux. Unlike the transport processes of the PIN photodiode,where photons generate carriers in a depletion region and are subject tohigh drift fields, carrier collection in the PN dotted photodiodedepends on much slower non-drift diffusion. For example, it might takeapproximately 1 microsecond for carriers to diffuse across 50 microns ofa non-depleted semiconductor region but less than 1 nanosecond to driftacross 50 microns in a depletion region of a PIN photodiode.

In accordance with one aspect of the present invention, the carrierdiffusion speed in the PN dotted photodiode can be significantlyimproved by fabricating the diode with substrate doping gradients havingcertain characteristics. One characteristic is that the majority dopinggradient in the substrate monotonically increases substantiallycontinuously from a lowest doping level adjacent to the dot PN junctionto a maximum doping level at points horizontal to the adjacent dot, atone-half of the distance to the adjacent dot, and vertical to the lightabsorption depth.

The increase in operational speed in the PN dotted photodiode accordingto this aspect of the present invention is caused by a field gradientproduced by the imbalance in majority carriers as they diffuse fromareas of high doping levels to those of lower doping levels. Since themajority carriers diffuse away from their associated acceptor or donordopant atoms, there is a net charge imbalance that creates an internalelectric field. This internal field gradient increases with dopinggradient. Consequently, an increase in the doping gradient produces acorresponding increase in minority carrier diffusion speed.

For data communications that do not need the increased bandwidth madepossible by the above described doping gradient technique, the currentcollecting diffusion dots can be placed further apart so as to providebandwidth that is sufficient for the application. By doing this, thecapacitance of the photodiode is reduced, which allows a decrease inreceiver input noise, improves the signal to noise ratio, and increaseseffective sensitivity. Also, placing the collecting dots of thedistributed photodiode structure further apart reduces the light maskingeffects caused by the inter-connecting metal used to connect the dots toform an electrical contact, which improves the optical gain of theresulting photodiode.

FIGS. 12, 13 and 14 are partial cross-sectional diagrams illustratingdopant concentrations in various embodiments of this aspect of thepresent invention. In the embodiments shown in FIGS. 12, 13 and 14, thesubstrate is shown as P type with N type collecting dots. However, thediode and techniques described can equally be applied to oppositepolarity substrate and collecting dots. However, it is important to notethat for the same dopant levels, a P substrate, as shown, tends toresult in faster performance than a comparable N substrate because theminority electron carriers naturally diffuse faster than minority holecarriers.

In the embodiment of a photodiode 1250 shown in FIG. 12, dopantgradients 1256A, 1256B and 1256C result from top down diffusions fromthe collection dots 1252A, 1252B and 1252C. The substrate dopinggradients on the surface 1251 of the dotted photodiode 1250 can beproduced using standard silicon processing techniques by diffusing adopant of the same polarity as the substrate at a site half way betweenthe collecting diffusion dots 1252A, 1252B and 1252C. In the example ofFIG. 12, a P substrate 1260 with N collecting dots 1252A, 1252B and1252C would have a P+ dopant diffused along a semi-circle radius from apoint half way between the N dots, as represented in cross-section by1254A, 1254B and 1254C. Other configurations for the plurality of P+diffusions 1254A, 1254B and 1254C may also be employed so long as the P+diffusions are disposed substantially between the N collection dots1252A, 1252B and 1252C. The electric field will be most negative wherethe P+ dopant is greatest, will decrease, and then gradually becomepositive as the P+ dopant concentration decreases. This field gradient,represented by areas 1256A, 1256B and 1256C in FIG. 12, will increasethe diffusion rate of electrons from the region of highest negativeelectric field to the region of least negative or most positive electricfield. Since the most negative field is at the furthest distance fromthe N dots 1252A, 1252B and 1252C, this has the effect of herding orincreasing the diffusion rate of the photon generated electrons towardthe collecting N dots 1252A, 1252B and 1252C.

In addition to creating a desirable field gradient, the surfacediffusion of the same polarity as the substrate helps prevent surfacepolarity inversions that can degrade the performance of the dottedphotodiode. These surface inversions act as isolated diodes whichcollect minority carrier charges, reducing the gain and bandwidth of thediode. These surface inversions are more likely to occur with a lightlydoped substrate, which is desirable for the fastest carrier diffusionspeeds. Thus, the surface diffusions 1254A, 1254B and 1254C indirectlyimprove speed by allowing use of extremely lightly doped substrates thatwould otherwise have problematic surface inversions.

FIG. 13 illustrates an embodiment of another photodiode 1300, accordingto this aspect of the present invention, that is formed by diffusingadditional substrate doping 1320 from the backside surface 1261 of thedotted photodiode in order to produce a vertical doping gradient 1322.In FIG. 13, the backside diffusion 1320 creates a dopant gradient 1322that results in a high dopant level at the light absorption depth 1324for the photodiode 1300 to a lower dopant level at the surface 1301 ofthe photodiode. This gradient may be achieved using a substrate 1330having a controllably thin thickness in order to minimize the dopantdiffusion distance and the resulting field gradient 1322. Very thinwafer and backside processing may be achieved through use of a handlingwafer in a process described in detail in U.S. Pat. No. 6,027,956 toPierre Irrisou, herein incorporated by reference in its entirety for allpurposes.

FIG. 14 shows yet another improved photodiode 1350, wherein the topsidegraded doping technique is used to diffuse dopants of the same polarityas the substrate into trenches 1352A, 1352B and 1352C between thecollecting diffusion dots 1252A, 1252B and 1252C. By diffusing thedopant in trenches 1352A, 1352B and 1352C, the doping gradient cangenerate fields deeper into the substrate 1330, speeding minoritycarrier collection for carrier pairs generated deep within the silicon.Further speed performance improvement is obtained when this trenchtechnique is combined with the bottom up vertical doping gradientillustrated in FIG. 13.

An alternative method that may be used to produce the vertical dopinggradient of this aspect of the present invention is to grow an undopedepitaxial layer over a suitably doped substrate. The epitaxial siliconlayer will pull dopant from the overgrown substrate in order to create adopant gradient of highest level some distance below the start of theepitaxial layer and then decreasing upwards to the top of the epitaxiallayer. In other words, the dopant will migrate from the substrate intothe epitaxial silicon layer during processing in such a manner as toproduce a dopant gradient that is strongest at the interface between thesubstrate and the epitaxial layer and gradually decreases as it reachestoward the top surface of the epitaxial layer. One of ordinary skill inthe art will readily recognize that many fabrication techniques may beapplied in order to obtain the doping gradient taught by the presentinvention.

The present invention is directed toward an improved process andstructure for a high speed PN photodiode having a distributed photodiodestructure. The PN dotted photodiode, according to the present invention,may be fabricated using low resolution lithography may have improvedperformance parameters of bandwidth, optical gain, and reducedcapacitance.

Having illustrated and described the principles of the present inventionin the context of the embodiments described above, it should be readilyapparent to those skilled in the art that the invention can be modifiedin arrangement and detail without departing from such principles.

I claim:
 1. A distributed photodiode, the photodiode comprising: asubstrate, the substrate being doped with a first dopant type and havingfirst and second planar surfaces; a first plurality of diffusions, thefirst plurality of diffusions being doped with a second dopant type andformed on the first planar surface of the substrate; a second pluralityof diffusions, the second plurality of diffusions being doped with thefirst dopant type and formed on the first planar surface of thesubstrate interposed the first plurality of diffusions; and a firstcontact having a first plurality of connective traces disposed on thefirst planar surface of the substrate and coupled to each of the firstplurality of diffusions.
 2. The distributed photodiode of claim 1,further including a backside diffusion formed on the second planarsurface of the substrate, the backside diffusion being doped with thefirst dopant type.
 3. The distributed photodiode of claim 2, furtherincluding a plurality of trenches formed in the first planar surface ofthe substrate, where each of the second plurality of diffusions isformed on a wall of a corresponding one of the plurality of trenches. 4.The distributed photodiode of claim 1, further including a plurality oftrenches formed in the first planar surface of the substrate, where eachof the second plurality of diffusions is formed on a wall of acorresponding one of the plurality of trenches.
 5. The distributedphotodiode of claim 1, wherein each of the second plurality ofdiffusions further comprises a substantially circular diffusion disposedaround a corresponding one of the first plurality of diffusions.
 6. Adistributed photodiode, the photodiode comprising: a substrate, thesubstrate being doped with a first dopant type and having first andsecond planar surfaces; a first plurality of diffusions, the firstplurality of diffusions being doped with a second dopant type and formedon the first planar surface of the substrate; a backside diffusionformed on the second planar surface of the substrate, the backsidediffusion being doped with the first dopant type; and a first contacthaving a first plurality of connective traces disposed on the firstplanar surface of the substrate and coupled to each of the firstplurality of diffusions.
 7. The distributed photodiode of claim 6,further including a second plurality of diffusions, the second pluralityof diffusions being doped with the first dopant type and formed on thefirst planar surface of the substrate interposed the first plurality ofdiffusions.
 8. The distributed photodiode of claim 7, further includinga plurality of trenches formed in the first planar surface of thesubstrate, where each of the second plurality of diffusions is formed ona wall of a corresponding one of the plurality of trenches.
 9. Thedistributed photodiode of claim 8, where each of the plurality oftrenches is substantially circular surrounding a corresponding one ofthe first plurality of diffusions.
 10. The distributed photodiode ofclaim 7, where each of the second plurality of diffusions issubstantially circular surrounding a corresponding one of the firstplurality of diffusions.